Soft switching flyback converter

ABSTRACT

Disclosed examples include synchronous rectifiers and flyback converters, integrated circuits and operating methods, in which a first switch is turned on to allow current to flow for a first time period in a first direction in a transformer primary winding responsive to a first switch voltage transitioning below a first threshold, and a second switch is turned on for a second time period after the first time period to transfer energy from a secondary transformer winding to drive a load. In the same converter cycle, the second switch is again turned on for a third time period in response to a second switch voltage transitioning below a second threshold at one of a series of troughs of a resonant ringing voltage waveform across the second switch, to cause current flow in a second direction in the primary winding to discharge a capacitance of the first switch to cause the first switch voltage to transition below the first threshold to initiate a subsequent converter cycle.

REFERENCE TO RELATED APPLICATION

Under 35 U.S.C. §119, this application claims priority to, and thebenefit of, U.S. provisional patent application No. 62/113,188, entitled“SOFT SWITCHING FLYBACK CONVERTER WITH PRIMARY CONTROL”, filed on Feb.6, 2015, the entirety of which application is hereby incorporated byreference.

TECHNICAL FIELD

The present disclosure relates generally to flyback converters, and moreparticularly to soft switching and synchronously rectified flybackconverters.

BACKGROUND

Synchronous rectifiers are used to perform DC-DC conversion in order todrive an output load, where a transformer is often used to construct aflyback converter with a secondary side switch to provide efficiencyadvantages over passive rectified flyback converters. In manyapplications, efficiency is a primary design goal, and it is desirableto reduce or mitigate switching losses as well as conduction losses inthe primary and secondary side switches. Soft switching or zero voltageswitching (ZVS) involves turning the primary and/or secondary sideswitches on when the voltage across the switch is low (preferably zero).Ideally, switching at zero volts minimizes switching loss, but this isdifficult due to drain-source capacitance of field effect transistor(FET) type switches. Conduction loss occurs while the switch is turnedon, and can be reduced by using larger switches, thereby reducing theon-state resistance (e.g., drain-source resistance RDSON for FETswitches). However, larger transistor dimensions leads to increase inthe switch capacitance, and thus simply increasing transistor size tomitigate conduction losses increases the switching losses, absent softswitching control. Furthermore, the ability to perform simple softswitching on the primary side switch of a synchronously rectifiedflyback converter is difficult over a wide range of input voltage andoutput voltage/current conditions. Certain conventional transition mode(TM) synchronous rectifier control schemes use valley control toregulate a converter output current or voltage, and the primary sideswitch is turned on at a local minima or “trough” of the resonantvoltage ring at the primary side switching node. However, the resonantvoltage oscillations do not approach zero volts even at the troughs,particularly for high input voltage conditions. Thus, true zero voltageswitching cannot be achieved across a wide range of operating conditionsfor conventional synchronously rectified flyback converters, andswitching losses can be substantial. At certain operating conditions,therefore, discontinuous mode (DM) switching operation must be used,which increases conduction losses and therefore reduces the converterefficiency. Moreover, hard switching (i.e., the inability to reliablyachieve true zero voltage switching) inhibits the ability to increaseswitch size for combating conduction losses, and leads to increasedcommon mode electromagnetic interference (EMI). Hard turn on of theprimary side switch can also cause resonant doubling of the voltage onthe secondary side rectifier, leading to increased synchronous rectifierblocking voltage as well as further increase to conduction losses causedby higher RDSON. Improved synchronous rectifier flyback converters andcontrol techniques are therefore desirable to mitigate capacitiveswitching losses to support increased power density and switchingfrequency without degraded efficiency.

SUMMARY

Example synchronous rectified flyback converters, integrated circuitsand operating methods are disclosed. A primary side first switch isturned on in response to its switch voltage transitioning below a firstthreshold to allow current to flow in a transformer primary winding fora first time period in a first direction. The first switch is turnedoff, and a secondary side switch is turned on for a second time periodto transfer energy from the transformer secondary to drive a load. Thesecond switch is again turned on in the same converter cycle for a thirdtime period in response to a second switch voltage transitioning below asecond threshold at one of a series of troughs of a resonant ringingvoltage waveform across the second switch. Turning off the second switchcauses current to flow in a second direction in the primary winding todischarge the first switch capacitance to cause the first switch voltageto transition below the first threshold to initiate a subsequentconverter cycle. These techniques and circuitry facilitate reduction inswitching losses, while allowing use of larger switch sizes to combatconduction loss, and converter operating switching frequency cantherefore be increased. Transformer leakage can be mitigated in certainexamples by use of a two-switch flyback topology to return leakageenergy to the input rather than dissipating it in the clamp. In certainexamples, the second actuation of the second switch is undertaken at ornear a particular trough in a series of resonant ringing peaks andtroughs of the second switch voltage. In certain examples, moreover, acontrol circuit selects a particular one of the troughs at which toagain turn on the second switch for the third time period at leastpartially according to a converter output signal. The trough selectionon the secondary side control is used to implement frequency modulationto regulate the converter output in certain examples, to providesecondary side regulation. In certain examples, primary side regulationis used.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a synchronously rectified flybackconverter with a first switch coupled with a transformer primary windingand operated by a first control circuit, as well as a second switchcoupled between a transformer secondary winding and an output load foroperation according to control signals from a second control circuit toimplement multiple secondary side switch actuations in a given convertercycle.

FIG. 2 is a flow diagram illustrating a method of operating asynchronously rectified flyback converter.

FIG. 3 is a waveform diagram of signals in the converter of FIG. 1 inwhich the secondary site switch is pulsed a second time in a givenconverter cycle at or near a second resonant trough of the second switchvoltage.

FIG. 4 is a waveform diagram of signals in the converter of FIG. 1, inwhich the secondary side switch is pulsed a second time in a givenconverter cycle at or near a fourth resonant trough of the second switchvoltage.

FIG. 5 is a schematic diagram of a second example synchronouslyrectified flyback converter with first and second primary side switches.

DETAILED DESCRIPTION

In the drawings, like reference numerals refer to like elementsthroughout, and the various features are not necessarily drawn to scale.In the following discussion and in the claims, the terms “including”,“includes”, “having”, “has”, “with”, or variants thereof are intended tobe inclusive in a manner similar to the term “comprising”, and thusshould be interpreted to mean “including, but not limited to . . . . ”Also, the term “couple” or “couples” is intended to include indirect ordirect electrical connection or combinations thereof. For example, if afirst device couples to or is coupled with a second device, thatconnection may be through a direct electrical connection, or through anindirect electrical connection via one or more intervening devices andconnections.

FIG. 1 shows a synchronously rectified flyback converter system 100,including a transformer 104, a primary side or first switch S1, and asecondary side or second switch S2 to convert input power from a DCvoltage source 102 to drive a load 125. The first switch S1 is operatedby a first switching control signal SC1 provided by a first controlcircuit 114, and the second switch S2 is operated according to a secondswitching control signal SC2 from a second control circuit 130. In oneexample, the switches S1 and S2 and the control circuits 114 and 130 areprovided in an integrated circuit (IC) 101 with terminals or pins orother suitable connections for receiving an input voltage VIN from anexternal DC source 102, one or more ground connections, connections toan external transformer 104, and to provide an output voltage VO to anexternal load 125. In other examples, the transformer 104 can beincluded within the IC 101. In other examples, a controller IC 101 canbe used, with suitable pins for connection to control external first andsecond switches S1 and S2. The illustrated examples include N-channelfield effect transistor (FET) switches S1 and S2. Other types ofswitches can be used, including without limitation P-channel FETs,bipolar transistors (P or N-type), insulated gate bipolar transistors(IGBTs), or the like, or combinations thereof, with the respectivecontrol circuits 114 and 130 providing suitable switching controlsignals SC1 and SC2 in order to actuate the switches S1 and S2 accordingto the principles of the present disclosure.

The transformer 104 includes a primary winding 108 and a secondarywinding 122, which are magnetically coupled with one another, forexample, by being wound at least partially around a common corestructure (not shown). The primary winding 108 includes a first end 106to receive the input voltage signal VIN, and a second end 110 connectedto the first switch S1. The first switch S1 is coupled between thesecond end 110 of the primary winding 108 and a first ground connectionor constant voltage node 112 (labeled GND1 in the drawing). The switchS1 includes a first terminal (e.g., a drain D) coupled with the secondend 110 of the primary winding 108, a second terminal (e.g., a source S)coupled with the first constant voltage node 112, and a first controlterminal (e.g., gate G) to receive the first switching control signalSC1. As also shown in FIG. 1, the switch S1 includes a body diode havingan anode connected to the source terminal, and a cathode connected tothe drain terminal, as well as a drain-source switch capacitanceschematically shown as a first switch capacitance CS1.

In operation, the first switch S1 is placed in an on state or conditionto selectively allow a first switch current IS1 to flow between thefirst and second terminals D, S when the first switching control signalSC1 is in a first state (e.g., HIGH for an N-channel FET S1). In thiscondition, a current ILm+IS1 flows from the input source 102 into thefirst end 106, including the current IS1 flowing through the firstswitch S1, and current ILm associated with a magnetizing inductance Lmof the primary winding 108. When the first switching control signal SC1is in a different second state (e.g., LOW), the first switch S1 is in anoff state or condition that prevents current flow between the first andsecond terminals D and S.

The secondary winding 122 of the transformer 104 includes a first end124 to provide an output voltage signal VO to drive the load 125, and asecond end 126 coupled with the second switch S2. The switch S2 includesa first terminal (e.g., drain D) coupled with the second end 126 of thesecondary winding 122, and a second terminal (e.g., source S) coupledwith a second constant voltage node 128, labeled GND2. In addition, theswitch S2 includes a second control terminal (e.g., gate G) coupled toreceive the second switching control signal SC2 from the second controlcircuit 130. As further shown in FIG. 1, the second switch S2 includes abody diode and a second switch capacitance shown as CS2. The secondswitch S2 operates in an on state or condition to allow a second switchcurrent IS2 to flow between its source and drain terminals D and S(e.g., between the second end 126 of the winding 122 and the secondconstant voltage node 128) when the second switching control signal SC2is in a first state (e.g., HIGH for a P-channel FET S2). In thisconfiguration, output current IO flows between the first end 124 of thesecondary winding 122 and the output load 125. S2 operates in an offstate or condition to prevent current flow between its source and drainterminals D and S when the signal SC2 is in a different second state(e.g., LOW). Although the illustrated example includes the secondaryside switch S2 in the lower circuit branch between the lower end 126 ofthe secondary winding 122 and the second constant voltage node GND2,other examples are possible in which the secondary side switch S2 isconnected between the upper end 124 and the load 125, with the lowersecondary winding end 126 coupled with GND2.

In operation, when the first and second switches S1 and S2 are turnedoff, a resonant isolation or ringing occurs in the second switch voltageof S2 (e.g., drain-source voltage VDS2 relative to GND2) due to thecharge transfer between the switch capacitance CS2, the inductance ofthe secondary winding 122, and capacitance of an output capacitor COconnected across the output terminals. This resonant ringing conditionresults in peaks and troughs in the voltage VDS2, where the troughsgenerally reach zero volts. Also, resonant ringing occurs in thisswitching condition in the voltage VDS1 across the primary side firstswitch S1, where the associated resonant peaks generally reach the inputvoltage level VIN, and the troughs generally do not reach zero. Turningthe first switch S1 on to begin a converter cycle at the troughs of thefirst switch voltage VDS1 can slightly reduce hard switching losses inS1, but this technique cannot provide a complete solution to switchingloss problems, and the situation worsens with increased input voltagelevels.

Referring also to FIGS. 2-4, the first and second control circuits 114and 130 in the disclosed examples advantageously provide for controlleddischarging of the first switch capacitance CS1 (and also the secondswitch capacitance CS2) by advanced switching control in which thesecond switch S2 is turned on prior to initiation of the primary sideswitching. FIG. 2 shows a process or method 200 for operating asynchronous rectifier, such as the flyback converter 100 of FIG. 1. Theprocess 200 is described below in connection with operation of the firstand second control circuits 114 and 130 in the IC 101 of FIG. 1.However, the methods of the present disclosure can be used in othersynchronous rectifier circuit configurations.

FIG. 3 illustrates example signal waveforms in the converter of FIG. 1,in which the secondary side switch S2 is pulsed a second time in a givenconverter cycle at or near a second resonant trough (e.g., local minima)of the second switch voltage VDS2. FIG. 4 shows an example where S2 ispulsed a second time at or near a fourth resonant trough in VDS2. Thewaveform diagrams include a graph 300 in FIG. 3 and a graph 400 in FIG.4 showing magnetizing current curves 302 (Thm) representing the currentassociated with the magnetizing inductance Lm of the primary winding108, as well as graphs 310 (FIG. 3) and 410 (FIG. 4) illustrating thefirst switch current curves 312 (IS1) flowing in the first switch S1.FIGS. 3 and 4 further provide graphs 320 and 420 respectivelyillustrating current curves 322 (IS2) flowing through the second switchS2. Graphs 330 and 430 in FIGS. 3 and 4 respectively show second switchvoltage curves 332 (VDS2) representing the drain-source voltage acrossthe second switch S2, and graphs 340 and 440 respectively illustratefirst switch voltage curves 342 (VDS1) representing the voltage acrossS1. In addition, graphs 350 and 450 illustrate first switching controlsignal curves 352 representing the gate-source control voltages VGS1provided by the first control circuit 114 to operate S1 (e.g., switchingcontrol signal SC1). The graphs 360 and 460 in FIGS. 3 and 4respectively show the second switching control signal curves 362 for thegate-source voltage VGS2 representing the second switching controlsignal SC2 provided by the second control circuit 130 to operate S2.

FIG. 2 shows the converter operation for a given one of a series ofconverter cycles. The process 200 begins at 202, where the first controlcircuit 114 provides the first switching control signal SC1 (high goingtransition in the curve 352 in graphs 350 and 450 in FIGS. 3 and 4) toturn on S1 for a non-zero first time period shown as period T1 in anexample converter cycle 301 in FIGS. 3 and 4. Turning on S1 allowscurrent IS1 to flow in a first (e.g., downward) direction in the primarywinding 108, indicated by the rising current flow in the ILm and IS1curves 302 and 312 in FIGS. 3 and 4. The first control circuit 114 turnson S1 in response to the first switch voltage VDS1 transitioning below afirst threshold VTH1, shown in the graphs 340 and 440 of FIGS. 3 and 4,respectively. Ideally, the first control circuit 114 turns on S1 whenthe voltage VDS1 is at or near zero, and thus a small threshold voltageVTH1 facilitates zero voltage or near-zero voltage switching. The VDS1curve 342 includes a falling edge 346 indicating the transition in VDS1below the threshold VTH1 at the beginning of the first time period T1A(FIG. 3). The second control circuit 130 turns off S2 via a rising edgein the SC2 control signal (curve 362), and the corresponding secondswitch voltage VDS2 oscillates around the output voltage level VO (curve332) during the first time period T1A.

As shown in FIG. 1, the first control circuit 114 in one exampleincludes a zero voltage switching (ZVS) circuit 118, which can include acomparator or other suitable circuitry to compare the switch nodevoltage VDS1 with a first threshold voltage VTH1 from a voltage source119. As schematically illustrated, the ZVS circuit 118 provides a signalto a first driver circuit 116, which in turn provides the firstswitching control signal SC1 to the gate control terminal of S1. In oneexample, in response to the sensed VDS1 signal transitioning below theVTH1 signal, the driver circuit 116 brings the SC1 signal high in orderto turn on the switch S1.

In certain examples, the first control circuit 114 also includes aclosed-loop (C-L) control circuitry 120 which controls the first timeperiod or duration T1 (T1A in FIG. 3, T1B in FIG. 4) during which thefirst switch S1 remains in the on state, based in whole or in part on adesired output level represented by a setpoint signal SP (e.g., adesired output current, voltage, power, etc.). The closed-loop controlcircuit 120 in one example can include one or more error amplifiers (notshown) generating an error signal to set the on-time T1 for S1 based oncomparison of the setpoint signal SP with one or more feedback signalsor values, such as the output current IO, the output voltage VO, etc.Where such primary-side regulation is implemented by the first controlcircuit 114, moreover, the IC 101 may include one or more isolationcircuits 142, for example, to provide an isolated feedback signal to thefirst control circuit 114 (relative to GND1) based on a signal sensed onthe secondary side of the transformer 104 (relative to GND2). The firstcontrol circuit 114 determines the first time period T1 according tosuch closed-loop control circuitry 120 in one example. In otherexamples, the first control circuit 114 implements a constant on time(COT) control scheme, where T1 is a generally constant value.

At 204 in FIG. 2, the first control circuit 114 turns off the switch S1.This causes the first switch voltage VDS1 to rise and oscillate for atime around the input voltage level VIN, shown in curve 342 in FIG. 3.This action reduces the first switch current IS1 to zero as shown in thecurve 312.

At 206 in FIG. 2, the second control circuit 130 turns on the secondaryswitch S2, causing the magnetizing inductance current Thm to beginramping down as shown in the curve 302, and the second switch currentIS2 starts to ramp down toward zero as shown in the curve 322. In oneexample, the second control circuit 130 includes a second driver circuit132 which provides a high gate control signal SC2 in order to turn onthe switch S2 at 206 in FIG. 2. The control circuit 130 maintains theswitch S2 in the initial on state for a non-zero second time period T2following the first time period T1 in the converter cycle 301, shown asT2A in FIG. 3 and T2B in FIG. 4. The second time period or duration T2can be a constant time in certain control schemes, for example, wherethe first control circuit 114 regulates the output condition of theconverter 100 by adjusting the on-time T1 of the first or primary sideswitch S1. In other examples, the second control circuit 130 can adjustthe second time period T2 according to one or more output conditions inorder to provide closed-loop regulation.

The second control circuit 130 turns the second switch S2 off at 208 inFIG. 2 at the end of the second time period T2. This ends the ramp downin the second switch current IS2, which levels off at zero. Turning offS2 causes resonant oscillation in the second switch voltage VDS2, shownin the indicated regions 334 and the curve 332 through operation of aresonant circuit formed by the inductance of the secondary winding 122and the capacitances CS2 and CO on the secondary side of the converter100. Turning off S2 while S1 remains off also causes correspondingresonant voltage swings in the primary side switch voltage VDS1 based onthe capacitance CS1 and the primary winding inductance, shown as peaksand troughs in the curve 342 in the circled regions 344 in FIG. 3. Thetrough of the resonant ring touches zero for any combination ofconverter input and output conditions (e.g., independent of VIN, VO,IO).

As seen in FIG. 1, one example of the second control circuit 130includes a logic circuit 134, a comparator or error amplifier 136, and asecond threshold voltage source 138 providing a second threshold voltagesignal VTH2 as an input to the comparator 136. The other comparatorinput in this example is connected to receive a secondary-side switchingnode signal representing the second switch voltage VDS2. The comparator136 provides an output signal to the logic circuit 134. The logiccircuit 130 in one example operates to provide first and second pulsesignals SC2 to turn the second switch S2 on and off twice in eachconverter cycle 301. In addition, the control circuit 130 in certainexamples also includes closed-loop (C-L) control circuitry 140. Theclosed-loop control circuitry 140 receives the output signal of thecomparator 136 and provides one or more signals to the logic circuit 134in order to implement a closed-loop regulation scheme to regulate one ormore output conditions of the converter 100 based on a setpoint signalor value SP and one or more feedback signals or values (e.g., IO, VO,etc.).

At 210 in FIG. 2, the second control circuit 130 monitors the resonantlyoscillating second switch voltage VDS2 (e.g., via the comparator 136).At 212, the second control circuit 130 again turns on S2 at 212 (thesecond actuation of S2 in a given converter cycle) at or near aparticular trough in VDS2 in response to VDS2 transitioning to or belowthe second threshold VTH2. The second control circuit 140 maintains thesecond switch S2 in the on state for a non-zero third time period T3 inthe converter cycle, shown as T3A in FIG. 3 and T3B in FIG. 4, and turnsoff S2 at 214 in FIG. 2. This second or supplemental actuation of S2 ina given converter cycle provides a slight reverse direction secondaryside current flow IS2 indicated in the circled regions 324 and the curve322, resulting in full or at least partial discharging of the secondswitch capacitance CS2. Turning S2 on also causes current flow IS1 inthe primary winding 108 in a second direction (e.g., upward in FIG. 1),shown as negative current transitions in the circled regions 314 in thecurve 312. The reversed primary side current flow IS1 fully or at leastpartially discharges the first switch capacitance CS1. Thesecondary-side initiated discharging of the CS1, in turn, decreases thefirst switch voltage VDS1 below the first threshold VTH1 at 216 in FIG.2, and thus causes the first control circuit 114 to initiate the next orsubsequent converter cycle 301.

The duration T3 during which the second switch S2 remains in the onstate can be adjusted by the second control circuit 130 in certainexamples. In one implementation, the second control circuit 130 monitorsVDS2 to detect a certain amount of discharge of CS2 in order todetermine when S2 is again to be turned off, thereby selectivelyadjusting the third time period T3. In other examples, T3 is apredetermined duration implemented by the second control circuit 130,and set to a value suitable to ensure sufficient discharge of theprimary-side first switch capacitance CS1 in order to trigger thebeginning of a subsequent converter cycle by the first control circuit114.

The second control circuit 130 in certain examples provides the secondsignal SC2 to turn on the second switch S2 for the third time period T3in response to the second switch voltage VDS2 transitioning below thesecond threshold VTH2 at or near a particular one of the troughs. Forinstance, the logic circuit 134 and/or the closed-loop control circuit140 can be configured to select a given trough in the second switchvoltage waveform resonant oscillations (e.g., curve 332 in graphs 330and 430 of FIGS. 3 and 4) at which the driver circuit 132 will betriggered. FIG. 3 shows one example in which the second control circuit130 uses the comparator 136 to monitor the peaks and troughs of theresonant ringing 334 in the VDS2 signal waveform 332. The logic circuit134 can include a counter or other suitable circuitry to select aparticular trough at which the switch S2 is turned on. In one example,the second control circuit 130 turns off the second switch S2 at 214 inFIG. 2 after a predetermined or fixed third time period T3 based on apredetermined number of rings in the voltage VDS2.

Comparing FIGS. 3 and 4, the second control circuit implements closedloop control using the circuitry 140 to selectively adjust the number oftroughs in VDS2 before S2 is turned on according to one or more feedbacksignals or values. This approach modifies the overall converter controlcycle or period T (e.g., TA in FIG. 3, TB in FIG. 4) and this changesthe switching frequency (1/T) to implement frequency modulated (FM)output regulation for the converter 100. As shown in FIGS. 3 and 4, forinstance, initiating the second switching of S2 at the second trough(FIG. 3) provides a first converter cycle duration TA, and waiting untilthe fourth trough to turn on S2 (FIG. 4) provides a longer convertercycle duration TB, thus decreasing the switching frequency. In certainexamples, the second control circuit 130 selects the particular troughat which to again turn on S2 at least partially according to a converteroutput signal VO, IO. This control capability can be used to providesecondary-side regulation of one or more converter output signals, suchas the output voltage VO, the output current IO, output power, or thelike. For example, the second control circuit 130 can select theparticular trough at which to again turn on S2 and thus set theconverter cycle duration T in order to selectively adjust the converterswitching frequency and the corresponding converter cycle duration T ofthe converter cycle 301 to at least partially regulate the converteroutput signal according to the setpoint signal SP. The second controlcircuit 130 in one example directly senses one or more converter outputsignals (e.g., VO and/or IO). In certain examples, the second controlcircuit 130 computes the converter output signal or signals according toone or more sensed conditions (e.g., VDS2, IS2, etc.) on the secondaryside of the transformer 104.

Using the above techniques, disclosed examples provide control ofconverter power transfer using a variety of different approaches,including magnetizing current amplitude modulation (AM) via primary-sideregulation in the first control circuit 114. In another example,frequency modulation (FM) can be performed using primary orsecondary-side regulation, in which the switching frequency may bereduced from its natural maximum (e.g., which occurs at Transition Mode)to lower values by starting a new switching cycle at a later trough ofthe voltage VDS2. Moreover, the use of the supplemental or auxiliaryactuation of S2 in each converter cycle provides enhanced mitigation orelimination of switching losses, and thus facilitates reduction of thefrequency modulation adjustment range needed to effectively reduce theconverter efficiency compared with conventionally controlled flybackconverters. The described examples also facilitate soft switching inboth converter switches S1 and S2, and recovery of the energy in theswitching node capacitances CS1 and CS2. In this manner, the efficiencybenefits of synchronous rectification are supplemented with reducedconduction loss in the clamp switch S1 and the transformer primary 108without addition of any new components, while allowing AM and FMmodulation for primary or secondary-side regulation of the outputconditions, and high efficiency even at high switching frequencies.

Also, the ability to achieve true or near ZVS switching at anyinput/output voltages and load combination facilitates maintainingtransition mode (TM) operation at much lower output loads than for aconventional flyback converter. This facilitates higher converterefficiency, lower acoustical noise and higher sampling frequency atlight load conditions. The amplitude of the resonant ring decays withtime, and eventually, the drain voltage of S1 settles at VIN and thedrain voltage of S2 settles at the output voltage level VO. Turning onS2 at this point in order to initiate the conduction of S1 significantlylowers the energy loss due to the forcible charge of the drain to commoncapacitances CS1 and CS2 compared with a conventional quasi-resonantconverter turning on at the same point because the voltage change on thecapacitors is considerably larger in the conventional case. Moreover,recycling of the energy stored in the capacitances CS1 and CS2 by thesecond actuation of S2 in the converter cycle facilitates oversizing S1and/or S2 to reduce conduction losses without the penalty of increasedloss due to the charging/discharging of these capacitances. OversizingS1, in turn, improves the self-snubbing when S1 is turned off, therebyreducing the turn off switching loss, and soft switching when S1 isturned on reduces common mode EMI.

Referring now to FIG. 5, disclosed examples also facilitate recovery oftransformer leakage energy, and thus allow significant increase of theswitching frequency. This loss can be reduced by using a two-switchflyback topology to return the leakage to the input instead ofdissipating it in the clamp. FIG. 5 shows a second example synchronouslyrectified flyback converter 100 using this two-switch flyback approach.The converter 100 in FIG. 5 includes first and second primary sideswitches S1 and S3, in addition to the secondary switch S2. In thisexample, S1 and S2 operate according to signals SC1 and SC2 from thefirst and second control circuits 114 and 130 as generally describedabove. The third switch S3 includes a first terminal (e.g., drain D)coupled to receive the input voltage signal VIN, a second terminal(e.g., source S) coupled with the first end 106 of the primary winding108, and a third control terminal (e.g., gate G) coupled to receive athird switching control signal SC3 from the first control circuit 114.In one example, the driver circuit 116 of the first control circuit 116provides the first and third simultaneous or concurrent switchingcontrol signals SC1 and SC3 in unison. In this manner, the first controlcircuit 114 provides the third switching control signal SC3 to turn onS3 when S1 is also on, and to turn off S3 when the first switch S1 turnsoff. A first diode D1 is coupled between the second end 110 of theprimary winding 108 and the input voltage VIN, and a second diode D2 iscoupled between the first end 106 of the primary winding 108 and thefirst constant voltage node 112 (GND1), so diodes D1 and D2 clamp theprimary winding 108 to the input voltage 102, thereby allowing recoveryof the leakage energy.

The above examples are merely illustrative of several possibleembodiments of various aspects of the present disclosure, whereinequivalent alterations and/or modifications will occur to others skilledin the art upon reading and understanding this specification and theannexed drawings. Modifications are possible in the describedembodiments, and other embodiments are possible, within the scope of theclaims.

The following is claimed:
 1. A flyback converter, comprising: a transformer, including: a primary winding, including a first end to receive an input voltage signal, and a second end, and a secondary winding, including a first end to provide an output voltage signal, and a second end; a first switch, including a first terminal coupled with the second end of the primary winding, a second terminal coupled with a first constant voltage node, and a first control terminal to receive a first switching control signal; a second switch, including a first terminal coupled with the second end of the secondary winding, a second terminal coupled with a second constant voltage node, and a second control terminal to receive a second switching control signal; a first control circuit to provide the first switching control signal to turn on the first switch for a non-zero first time period in a converter cycle to allow current to flow in a first direction in the primary winding in response to a first switch voltage across the first switch transitioning below a first threshold; and a second control circuit to provide the second switching control signal to turn on the second switch for a non-zero second time period following the first time period in the converter cycle, the second control circuit operative to provide the second switching control signal to again turn on the second switch for a non-zero third time period in the converter cycle in response to a second switch voltage across the second switch transitioning below a second threshold to cause current flow in a second direction in the primary winding to at least partially discharge a capacitance of the first switch to cause the first switch voltage to transition below the first threshold to initiate a subsequent converter cycle.
 2. The flyback converter of claim 1, wherein the second switch voltage undergoes resonant ringing including a series of peaks and troughs while the second switch is off; and wherein the second control circuit is operative to provide the second switching control signal to turn on the second switch for the third time period in response to the second switch voltage transitioning below the second threshold at or near a particular one of the troughs.
 3. The flyback converter of claim 2, wherein the second control circuit selects the particular one of the troughs at which to again turn on the second switch for the third time period at least partially according to a converter output signal.
 4. The flyback converter of claim 3, wherein the second control circuit selects the particular one of the troughs at which to again turn on the second switch for the third time period to selectively adjust a switching frequency and a duration of the converter cycle to at least partially regulate the converter output signal according to a setpoint signal.
 5. The flyback converter of claim 4, wherein the second control circuit directly senses the converter output signal.
 6. The flyback converter of claim 4, wherein the second control circuit computes the converter output signal according to one or more sensed conditions on the secondary side of the transformer.
 7. The flyback converter of claim 1, wherein the first control circuit selectively adjusts the first time period to regulate a converter output signal.
 8. The flyback converter of claim 1, comprising: a third switch, including a first terminal coupled to receive the input voltage signal, a second terminal coupled with the first end of the primary winding, and a third control terminal to receive a third switching control signal; a first diode, including a first anode connected to the second end of the primary winding, and a first cathode connected to the first terminal of the third switch; and a second diode, including a second anode connected to the first constant voltage node, and a second cathode connected to the first end of the primary winding; wherein the first control circuit is operative to provide the third switching control signal to turn on the third switch when the first switch is on, and to turn off the third switch when the first switch is off.
 9. The flyback converter of claim 8, wherein the second switch voltage undergoes resonant ringing including a series of peaks and troughs while the second switch is off; and wherein the second control circuit is operative to provide the second switching control signal to turn on the second switch for the third time period in response to the second switch voltage across the second switch transitioning below the second threshold at or near a particular one of the troughs.
 10. The flyback converter of claim 9, wherein the second control circuit selects the particular one of the plurality of troughs at which to again turn on the second switch for the third time period at least partially according to a converter output signal.
 11. An integrated circuit (IC) to operate a flyback converter, comprising: a first switch, including a first terminal to couple with a primary winding of a transformer, a second terminal to couple with a first constant voltage node, and a first control terminal to receive a first switching control signal; a second switch, including a first terminal to couple with a secondary winding of the transformer, a second terminal to couple with a second constant voltage node, and a second control terminal to receive a second switching control signal; a first control circuit to provide the first switching control signal to turn on the first switch for a non-zero first time period in a converter cycle to allow current to flow in a first direction in the primary winding in response to a first switch voltage across the first switch transitioning below a first threshold; and a second control circuit to provide the second switching control signal to turn on the second switch for a non-zero second time period following the first time period in the converter cycle, the second control circuit operative to provide the second switching control signal to again turn on the second switch for a non-zero third time period in the converter cycle in response to a second switch voltage across the second switch transitioning below a second threshold to cause current flow in a second direction in the primary winding to at least partially discharge a capacitance of the first switch to cause the first switch voltage to transition below the first threshold to initiate a subsequent converter cycle.
 12. The IC of claim 11, wherein the second switch voltage undergoes resonant ringing including a series of peaks and troughs while the second switch is off; and wherein the second control circuit is operative to provide the second switching control signal to turn on the second switch for the third time period in response to the second switch voltage transitioning below the second threshold at or near a particular one of the troughs.
 13. The IC of claim 12, wherein the second control circuit selects the particular one of the troughs at which to again turn on the second switch for the third time period at least partially according to a converter output signal.
 14. The IC of claim 13, wherein the second control circuit selects the particular one of the troughs at which to again turn on the second switch for the third time period to selectively adjust a switching frequency and a duration of the converter cycle to at least partially regulate the converter output signal according to a setpoint signal.
 15. The IC of claim 11, wherein the first control circuit selectively adjusts the first time period to regulate a converter output signal.
 16. The IC of claim 11, comprising: a third switch, including a first terminal coupled to receive an input voltage signal, a second terminal coupled with a first end of the primary winding, and a third control terminal to receive a third switching control signal; a first diode, including a first anode connected to a second end of the primary winding, and a first cathode connected to the first terminal of the third switch; and a second diode, including a second anode connected to the first constant voltage node, and a second cathode connected to the first end of the primary winding; wherein the first terminal of the first switch is coupled with the second end of the primary winding; and wherein the first control circuit is operative to provide the third switching control signal to turn on the third switch when the first switch is on, and to turn off the third switch when the first switch is off.
 17. A method of operating a flyback converter having a first switch coupled with a primary winding of a converter transformer and a second switch coupled with a secondary winding of the converter transformer, the method comprising, in each of a series of converter cycles: turning on the first switch for a non-zero first time period to allow current to flow in a first direction in the primary winding in response to a first switch voltage across the first switch transitioning below a first threshold; turning off the first switch after the first time period; turning on the second switch for a non-zero second time period after the first time period; turning off the second switch after the second time period; turning on the second switch again for a non-zero third time period in response to a second switch voltage across the second switch transitioning below a second threshold to cause current flow in a second direction in the primary winding to at least partially discharge a capacitance of the first switch to cause the first switch voltage to transition below the first threshold to initiate a subsequent converter cycle.
 18. The method of claim 17, wherein the second switch voltage undergoes resonant ringing including a series of peaks and troughs while the second switch is off; the method further comprising turning on the second switch for the third time period in response to the second switch voltage transitioning below the second threshold at or near a particular one of the troughs.
 19. The method of claim 18, further comprising selecting the particular one of the troughs at which to again turn on the second switch for the third time period at least partially according to a converter output signal.
 20. The method of claim 19, further comprising selecting the particular one of the troughs at which to again turn on the second switch for the third time period to selectively adjust a switching frequency and a duration of the converter cycle to at least partially regulate the converter output signal. 